Wide bandwidth, current sharing, MOSFET audio power amplifier with multiple feedback loops

ABSTRACT

The disclosure describes a phase-splitter/level shifter comprising a first MOS transistor has a drain coupled to the first output node and a source coupled to a feedback node through a source resistor and a gate. A second MOS transistor has a drain coupled to a second output node and a source coupled to the feedback node through a source resistor and a gate. A first operational amplifier has a non-inverting input coupled to a single-ended input node, and an inverting input coupled to a second reference current source, an output coupled to the gate of the first MOS transistor. A second operational amplifier has a non-inverting input coupled to a single-ended input node, an inverting input coupled to a first reference current source, and an output coupled to the gate of the second MOS transistor. A variable resistor is coupled between the source of the first and second MOS transistor.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a Continuation In Part of a Continuation applicationSer. No. 09/896,850, filed Jun. 28, 2001 and now U.S. Pat. No. 6,414,549which is a Continuation of U.S. patent application Ser. No. 09/415,039,filed Oct. 7, 1999 and now U.S. Pat. No. 6,268,770 which is aContinuation of U.S. patent application Ser. No. 09/118,195, filed Jul.17, 1998, now U.S. Pat. No. 6,144,256, which is a Continuation of U.S.patent application Ser. No. 08/774,537, filed Dec. 30, 1996, now U.S.Pat. No. 5,815,040.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention pertains to audio amplifiers. More particularly,the present invention pertains to wide bandwidth, current sharing,mosfet audio power amplifier with multiple feedback loops.

2. The Prior Art

MOSFET transistors have a number of advantages when used in audio poweroutput stages. The complementary push-pull source follower stage such asis typically employed in such applications is essentially a voltageoperated current output device with extremely high input impedance, thusrequiring very small drive currents in the micro-ampere range. Thesesmall drive currents are well within the output capability of low levellinear circuits such as monolithic operational amplifiers.

Such MOSFET transistors are extremely stable in the source followerconnection because they have a voltage gain of less than unity. Further,they are easily driven to the power supply rails using traditionalcapacitor bootstrap techniques, and they perform well in releasing fromthe supply rail during maximum voltage excursion since they havevirtually no internal carrier storage time, unlike bipolar devices whichhave a tendency to “stick” to the rail due to carrier storage.

Device capacitance presents little or no degradation of speed of MOSFETtransistors in the linear mode because only very small currents areneeded to modulate the gate voltage under dynamic conditions such asrequired in audio power amplifier applications. This feature isconsidered to contribute to the extremely low transient intermodulationdistortion performance of these devices.

While they have a certain amount of capacitance between the gate, sourceand drain, which must be charged and discharged by the gate drivemechanism, the impact on device bandwidth is small for two reasons.First, when driving the MOSFET devices in the linear region, only smallchanges in gate voltage are necessary to command large changes indrain/source current. Thus, only small currents are necessary to causeappropriate gate voltage changes. Second, bandwidth is virtuallyun-affected because the typical 100 ohm resistor commonly used as a gateisolation device and which forms a low pass filter with the devicecapacitance, typically has a pole at several megahertz, which is wellabove any recognized bandwidth of interest with regard to audioapplications.

One of the fundamental disadvantages of MOSFET devices is that theyexhibit a wide variation in gate threshold voltage among individualdevices having the same part number, and a temperature coefficient ofgate threshold that does not relate to bi-polar transistors, which makesit difficult to establish a quiescent bias point using traditionaltechniques.

In traditional class AB operation, which is defined as a conductionangle of 180 degrees for each transistor, provisions are made for theconduction period to overlap a few degrees to eliminate crossoverdistortion. Typically, large amounts of negative feedback are alsoemployed to minimize this and other forms of distortion. In establishedtopologies, the feedback connection is not sufficient to fully removethis error component from the audio output, which makes a certain amountof conduction overlap imperative. Audio power stages with very largeconduction overlap, such as class A amplifiers, are considered to havevery low audible distortion but achieve this at the expense of very highpower dissipation.

Large conduction overlap represents a power drain on the system, andgenerates heating of the output devices that must be addressed inheatsink design as well as device selection and sizing, and physicalcircuit layout. It would be advantageous to design a system that willprovide a stable overlap region which is high enough to preventcrossover distortion yet low enough to minimize power dissipation withno input signal applied. Quiescent level power dissipation generates anundesirable temperature rise in the transistor heatsink.

In MOSFET devices there is also a lack of correlation between initialthreshold voltage and linear transconductance, which makes themextremely difficult to match by selection of devices from a large batchof supposedly identical devices. This means that when devices areconnected in parallel, they have to be matched for at least twodifferent and un-related parameters: that of the transistor underquiescent conditions, and that of the transistor under load. This is acostly process, and in order to compensate for changes in matching ofdevice characteristics with component aging, rational design wouldrequire excessive component de-rating, further raising the cost of aproduct into which they are designed

Even with a single pair of complimentary output devices, quiescent biaspresents difficulties because bipolar devices, which are convenient touse in driver and voltage gain stages, exhibit a temperature coefficientof −2.2 mv. per degree centigrade as opposed to −5 to −7 mv per degreeC. for MOSFET devices. Topologies can be devised to address thesedifficulties, but with virtually all of them, including those presentedhere, there remains a wide variation in output impedance, or dampingfactor in the crossover region. This is a significant cause of crossoverdistortion, which is well understood to be a particularly audible, andthus undesirable, form of power amplifier distortion.

Further, even the driving circuitry itself contributes to the tendencyof audio amplifiers to exhibit wide variations in damping factor withchanges in power level and frequency. It would be desirable to provide afeedback technique that will alleviate this effect. Such a techniquewould desirably be applied to amplifiers with other types of transistorsin the output stage with equally advantageous effect.

Device manufacturers offer exhaustive descriptions of MOSFET parametersand their variations, as well as in-depth studies and discussions of thedevice characteristics and their implications. These are invaluable informulating an understanding of the requirements for an easily biasedand stable power output stage, but they do not suggest circuitry thatwill provide functional solutions for production designs.

Some commercially available audio products do successfully address theseproblems, although matching of components and/or sensitive circuitadjustments are not eliminated. Using a MOSFET as the biasing elementfor a single pair of output transistors is an effective technique, butit does not address the question of component variation in topologieswhere devices are connected in parallel. As a result, MOSFET poweramplifiers have been confined to the realm of modest output power, or toexpensive, hand built, esoteric audio products intended for a very smallsegment of the audio market. This leaves access to their many advantageseconomically out of reach for products intended to be sold atcompetitive prices.

It is thus an object of the present invention to provide a MOSFET audioamplifier circuit which overcomes some of the shortcomings of the priorart.

Another object of the present invention to provide a MOSFET audioamplifier circuit which is able to utilize the advantageous propertiesof MOSFET devices to as great an extent as possible.

A further object of the present invention to provide a MOSFET audioamplifier circuit which provides current sharing for output devices andhas a high bandwidth.

These and other objects and advantages of the present invention willbecome apparent from the disclosure herein.

BRIEF DESCRIPTION OF THE INVENTION

An audio amplifier according to the present invention drives a pluralityof paralleled current shared individual MOS output transistors. An audioinput is supplied to a voltage feedback amplifier stage having an audiosignal input. The voltage feedback amplifier stage drives a push-pullvoltage gain/phase splitter stage. A bias adjustment stage is drivenfrom the push-pull voltage gain/phase splitter stage. A current drivestage is driven from the bias adjustment stage. The current drive stagedrives an output stage comprising a plurality of paralleled currentshared individual MOS output transistors driving an output node. Anoutput load has a first terminal connected to the output node and asecond terminal connected to a fixed voltage potential.

The amplifier of the present invention includes up to three feedbackloops. A first voltage feedback loop comprises a voltage feedback stagehaving an input connected to a voltage divider driven from the firstterminal of the load and an output connected to a feedback input node inthe voltage feedback amplifier stage. A second voltage feedback loopcomprises a voltage feedback stage having an input connected to thefirst terminal of the load and an output connected to a feedback inputnode in the push-pull voltage gain/phase splitter stage. A thirdfeedback loop comprises a current feedback stage having an input inseries between the output node and the load and an output connected to afeedback input node in the voltage feedback amplifier stage. The currentfeedback connection works in conjunction with input stage to lowerdistortion and provide a relatively flat frequency response.

Current sharing of MOS output devices is made possible by the circuitconfiguration comprising an operational amplifier in the output stage.Inherent current limiting is provided due to resistor strings used tobias the inputs of the operational amplifier in the output stage. Theseresistor strings provide numerous possible modulation nodes.

The amplifier of the present invention has a high degree of temperaturestability due to employment of temperature stable references in thebiasing network of the operational amplifier in the output stage and inthe voltage gain/phase splitter stage. The configuration of the inputstage lowers the damping factor of the amplifier of the presentinvention.

BRIEF DESCRIPTION OF THE DRAWING FIGURES

FIG. 1 is a schematic diagram of a typical output stage of a MOSFETaudio amplifier.

FIG. 2 is a block diagram of a MOSFET output audio power amplifieraccording to a presently preferred embodiment of the invention.

FIGS. 3a, 3 b, and 3 c, are schematic diagrams of alternate embodimentsof a voltage feedback amplifier stage suitable for use in the presentinvention.

FIG. 4 is a schematic diagram illustrating the voltage divider formed bythe output impedance as a small resistance in series with the loadresistance.

FIG. 5 is a schematic diagram of a feedback loop employing adifferential sensing amplifier.

FIG. 6 is a schematic diagram of a bipolar voltage gain/phase splitterstage, suitable for use in the present invention.

FIG. 7a is a schematic diagram of a MOSFET voltage gain/phase splitterstage, suitable for use in the present invention.

FIG. 7b is a schematic diagram of an alternate MOSFET voltage gain/phasesplitter stage suitable for use in the present invention. FIG. 8 is aschematic diagram of the positive half of an output, or current gainstage for use in the amplifier of FIG. 2 FIGS. 9 through 15 areschematic diagrams illustrating numerous ways to introduce audiomodulating voltage inputs to the current gain stage circuit of FIG. 8.

FIG. 16 is a schematic diagram illustrating the driving of multipleMOSFET transistors connected in parallel, including the components thatmust be duplicated and those that are shared by all sections of thecircuit.

FIG. 17 is a graph showing speaker voice coil impedance vs. frequencyfor typical loudspeakers as measured in free air for a typical amplifiercircuit employing current feedback techniques.

FIG. 18 is a schematic diagram of an audio amplifier according to thepresent invention employing a circuit for developing a voltage which maybe applied to a current feedback amplifier.

FIGS. 19a and 19 b are schematic diagrams illustrating circuits forsupplying bias voltages to the amplifier circuits disclosed herein.

FIG. 20 is a schematic diagram of an alternate MOSFET voltage gain/phasesplitter stage suitable for use in the present invention.

FIG. 21 is a schematic diagram of an alternate MOSFET voltage gain/phasesplitter stage suitable for use in the present invention.

FIG. 22 is a schematic diagram of an alternate MOSFET voltage gain/phasesplitter stage for use in the present invention.

DETAILED DESCRIPTION OF A PREFERRED EMBODIMENT

Those of ordinary skill in the art will realize that the followingdescription of the present invention is illustrative only and not in anyway limiting. Other embodiments of the invention will readily suggestthemselves to such skilled persons.

In the disclosure presented herein, MOSFET devices disclosed hereininclude at least single transistor devices, such as MTP 12 N 20, MTP 12P 20, available from sources such as Motorola Semiconductor of PhoenixAriz., and single or multiple transistor functional equivalents thereof,whether now known or unknown.

FIG. 1 is a schematic diagram which shows a typical prior-art outputstage of a MOSFET audio power amplifier. The amplifier of FIG. 1 suffersfrom the aforementioned drawbacks relating to transistor matchingproblems.

Referring now to FIG. 2, a diagram of an amplifier 10 is presented whichcan employ MOSFET transistors in the voltage gain and current gainstages. The amplifier 10 of FIG. 2 employs unique modulating techniquesdisclosed herein which allow stable and easily adjusted quiescentbiasing.

The first stage of amplifier 10 comprises a voltage feedback amplifier12. The output of voltage feedback amplifier 12 is presented to voltagegain/phase splitter stage 14. Voltage gain/phase splitter stage 14drives current drive stage 16 through bias adjustment stage 18. Currentdrive stage 16 drives an output stage 20 comprising paralleledindividual output devices. Four individual MOSFET output devices 22, 24,26, and 28 are shown in FIG. 2, but persons of ordinary skill in the artwill recognize that the amplifier 10 of the present invention may employlarger numbers of paralleled output devices.

Referring now to FIGS. 3a, 3 b, and 3 c, alternate embodiments ofvoltage feedback amplifier stage 12 are shown in schematic diagram form.Turning first to FIG. 3a, the voltage feedback amplifier stage 12illustrated therein is formed around operational amplifier 30.Operational amplifier 30 may be one of many commonly used operationalamplifier circuits, but low noise operational amplifiers, such as theNE5532 operational amplifier available from Signetics of Sunnyvale,Calif. may be advantageously employed in the present invention.

When the voltage feedback amplifier of FIG. 3a is employed in thepresent invention, the audio input to amplifier 10 of the presentinvention is provided to input node 32, connected to the non-invertinginput of operational amplifier 30. The inverting input to operationalamplifier 30 is connected to the output of a voltage divider comprisingfeedback resistor 34 and shunt resistor 36. The values of feedbackresistor 34 and shunt resistor 36 are chosen such that the gain of theamplifier 10 is between about 10 and about 50.

Referring now to FIG. 3b, a schematic diagram of a presently preferredvoltage feedback amplifier for use in the present invention is shown.Audio input node 32 of amplifier 10 is AC coupled through a well-knownhigh-pass network including coupling capacitor 38 which may have a valueof, for example 10 μF, and resistor 40, which may have a value of forexample about 10-20 KΩ. Using these values for capacitor 38 and resistor40 results in a low-end frequency response of about 10 Hz for theamplifier 10.

A low-pass filter comprising series resistor 42 and shunt capacitor 44suppresses parasitic RF oscillation of amplifier 10 caused by strayradio frequency detection. The typical cutoff frequency of the low passfilter is about 40 KHz and typical values for the low-pass filter are 2KΩ for resistor 42 and 1000 pF for resistor 44.

The output of the low-pass filter is presented to the non-invertinginput of operational amplifier 46. The inverting input of operationalamplifier 46 is fed through feedback resistor 48 and bias resistor 50.Feedback resistor 48 is preferably a variable resistor, such as atrimpot, to allow setting of the gain structure of the operationalamplifier 46.

In the embodiment of FIG. 3b, the output of the voltage dividercomprising feedback resistor 34 and shunt resistor 36 is presented tooperational amplifier 52, configured as a well-known source follower. Athird operational amplifier 56 has its non-inverting input fed from theoutput of operational amplifier 46 and its inverting input fed from theoutput of source follower amplifier 52, through resistor 54. Feedbackresistor 58 sets the gain of operational amplifier 56, preferably fromabout 10 to about 100.

FIG. 3c is a schematic diagram of an alternate embodiment of a voltagefeedback amplifier which may be employed in the present invention. Froman examination of FIG. 3c, those of ordinary skill in the art willrecognize that the source follower configuration of operationalamplifier 52 has been replaced by a negative feedback configurationwhich provides additional gain to the feedback signal through the use ofresistors 60 and 62. As presently preferred, the gain of operationalamplifier 52 of FIG. 3c is set to about 10 by choosing the values ofresistors 60 and 62 to be, for example, about 1 KΩ and 10 KΩrespectively. Persons of ordinary skill in the art will recognize thatdifferent gain structures may be employed as warranted for differentapplications.

The voltage feedback amplifier stage 12 of the amplifier 10 of FIG. 2provides moderate gains to the input signal while applying large amountsof gain to the error component of the feedback signal. This has theeffect of raising the output damping factor. The current senseresistors, to be described later as part of the output stage biasing andcurrent control circuitry, have the disadvantage that they can degradethe damping factor of the amplifier.

Damping factor is defined as the inverse of output impedance. (Dampingfactor=1/Z out) Further, in cases where a current feedback system isutilized in the power amplifier, damping factor is grossly degraded bythe sense resistors normally found in this type of circuit. For example,the typical sense resistor used in output stages employing currentfeedback designs has a value of 0.1 ohm. This component is essentiallyplaced in series with the power amplifier output, with the result thatthe best case damping factor cannot be greater than 10 (1/0.1=10). Thisis considered to be quite poor for high quality audio equipment.

It is generally understood that feedback lowers the output impedance ofan audio amplifier. This is a function of a feedback voltage beingintroduced to the input stage of the multi-stage amplifier. Traditionalfeedback does lower output impedance considerably, but is far from fullyeffective. This is evidenced by the fact that there appear to be noaudio amplifiers in existence with damping factors approaching infinity,(corresponding to an ideal output impedance of zero), which would be thecase if traditional voltage feedback were fully effective. Therefore, itis possible to conclude that traditional feedback is not sufficient tocompletely remove voltage errors from the power amplifier output

Output impedance can be characterized as a small resistance in serieswith the load resistance. In this sense, a voltage divider is formed, asillustrated in FIG. 4. The output voltage of the amplifier may beconsidered as having been attenuated by this divider. Therefore, themagnitude of amplifier output impedance can be represented as a voltageerror which can be fed back to an error amplifier and divided by thegain of that amplifier. In traditional connections, gain is applied tothe error signal, but gain is also concurrently applied to the inputsignal as well. This is because the open loop gain of the erroramplifier is applied to both the input signal and the error signal.

In prior-art circuits, the feedback loop of A! is closed to the outputof the power amplifier, as in FIG.3a. In FIG. 3b, a connection is shownthat applies additional gain to the error signal while maintaining unitygain for the input signal. In the arrangement shown in FIG. 3b, theerror amplifier has a closed loop gain of unity for the input signalsince resistor 54 does not transmit current to the inverting input ofoperational amplifier 56. This is because the output of operationalamplifier 52 is driven with much the same voltage as the input signalapplied to the non-inverting input of amplifier 56.

At the same time, error components of the feedback signal, which can beconsidered as small variations superimposed on a replica of the inputsignal, will appear at the output of operational amplifier 52 of FIG.3b, are amplified with a gain of (−R34/R36)×Verror. Resistors 34 and 36are chosen such that the voltage divider divides the output voltage bythe same ratio as the gain applied by amplifier 14 of FIG. 2. Thisadditional gain, which is applied only to the error signal in thefeedback loop, results in additional lowering of the output impedance ofthe power amplifier. Even greater gains may be applied with the additionof the connection shown in FIG. 3c. This circuit compares the inputvoltage to the error voltage in two different stages and can apply apotential gain to the error signal that is many times greater than canbe applied by a traditional error amplifier.

Referring now to FIG. 5, in the case of a system using a current senseresistor, or in systems employing long speaker leads which themselvesappear as complex impedances with resistance, inductance andcapacitance, and also introduce errors, a differential sensing amplifier70 may be employed for feedback with good results.

In the mode illustrated in FIG. 5, separate leads 72 and 74 would beprovided to detect the error voltage remotely at the input terminals tothe speaker system 76, as in the traditional remote sense arrangementsemployed widely in applying DC power supplies. Operational amplifier 70is employed as a differential feedback signal amplifier, having a gainselected by resistors 78, 80, 82, and 84. The arrangement shown in FIG.5 reduces audible distortion in systems that employ long speaker leadssuch as in large concert sound installations or distributed soundsystems in large buildings. Those of ordinary skill in the art willrecognize that a phase inversion may be implemented in this stage byexchanging the connections of resistors 78 and 82.

A bipolar voltage gain/phase splitter stage, such as shown in FIG. 6, iswell suited to be used with the new circuits shown here. Circuitry suchas is depicted in FIG. 6 has long been in use and is well understood inthe art. Explanation of the details of the operation of such circuitryis omitted herein in order to avoid unnecessarily obfuscating thedisclosure of the present invention.

A schematic diagram of a MOSFET analogy of the bipolar voltagegain/phase splitter stage circuit of FIG. 6 is shown in FIG. 7a. Thevoltage gain/phase splitter stage circuit 90 of FIG. 7a includes activebiasing to promote stability as well as high speed. In the positive(top) half of the MOSFET voltage gain/phase splitter stage embodimentshown in FIG. 7a, an operational amplifier 92 drives a MOSFET 94 tocompare the voltage across resistor 96 with a reference voltage. In thenegative (bottom) half of the MOSFET voltage gain/phase splitter stageembodiment shown in FIG. 7a, an operational amplifier 98 drives a MOSFET100 to compare the voltage across resistor 102 with the referencevoltage. In each case, operational amplifiers (respectively shown atreference numerals 104 and 106), drive MOSFET devices (respectivelyshown at reference numerals 108 and 110).

A temperature stable reference 112 is preferably employed as referencevoltage source to guarantee that the currents in all devices will remainconstant, as disclosed herein. An example of such an adjustabletemperature-stable reference which may be employed in the presentinvention is LM385Z, available from National Semiconductor of SantaClara, Calif.

The supply voltage applied to the source resistors 114 and 116 may beset to between about ±40V and about ±80V, while the supply voltageapplied to operational amplifiers 92 and 98 may be between about ±10Vand about ±15V. Those of ordinary skill in the art will recognize thatthe supply voltage applied to the source resistors 114 and 116 willdetermine the maximum power output obtainable from amplifier 10.

Persons of ordinary skill in the art will recognize that the circuits ofFIGS. 5 and 6 may also be partially combined, employing the common-baseinput bipolar transistors of FIG. 6 with the MOSFET transistors 108 and110 to the right of the dashed line of FIG. 7a. Such skilled personswill also appreciate the presence of series resistors driving the gatesof MOSFET devices 94, 100, 108, and 110 and will set the values of theseresistors in order to damp potential oscillations of MOSFETS 94, 100,108, and 110 by introducing an RC frequency pole at the gates of theMOSFETS. Typical values for these resistors are in the range of about100 Ω to about 1 KΩ.

As an alternative, the MOSFET bias provided in this stage by operationalamplifiers 92 and 98, may be generated with the diode string of FIG. 6in series with a resistor as shown in FIG. 6.

In the embodiment depicted in FIG. 6, the diodes, which may be 1N4001signal diodes, available from many sources, will typically exhibit atemperature coefficient of 3000 ppm per ° C., while the resistors in theresistor string are considered to be relatively stable, (typically50ppm). The temperature-sensitive voltage drop of the diode stringrepresents a certain percentage of the total gate bias voltage, asdeveloped by the resistor string and typical ±15 volt system voltagesupply sources. With appropriate component selection, an example ofwhich is R=(1 to 2 volts)×Ibias, the present invention contemplates thata bias voltage can be generated that closely emulates the temperaturesensitivity of the MOSFET threshold voltage, which is typically in therange of 1000 to 1500 ppm/degree C.

The operational amplifier 92 will cause the MOSFET 94 to drive currentinto the resistor 96 until the two voltages across the inverting andnon-inverting inputs of operational amplifier 92 are equal, and thenstabilize. The current in the transistor 94 is therefore defined andstabilized. The resulting voltage across source resistor 118 is alsofully defined by the same current and the selected value of the resistor96.

The selected values of resistor 96, and its counterpart resistor 102 inthe negative half of the circuit, are made to set the voltage inresistors 118 and 120 to values that are required for the operation of apractical circuit. As shown in FIG. 21, further variable adjustment ofthis voltage may also be made through the introduction of an additionalresistance network, comprised of resistor 400 connected in series withresistor 401, and both connected in parallel with resistor 96 andresistor 102.

Practitioners of the art will recognize that in the disclosed circuit,the current flowing in MOSFET 94 and MOSFET 100 will be defined by thetotal of all parallel resistances connected between their respectivesource terminals. As previously discussed, these parallel resistances,by defining the current flowing in MOSFET 94 and MOSFET 100, willtherefore define the voltages across resistors 118 and 120. Thisparallel resistance path would typically be 10 to 100 times higher thanthe practical values of resistors 96 and 102. Potentiometer 401 wouldalso typically be 10 times smaller than resistor 400.

FIG. 21 also diagrams how the existing circuit can be connected to drivea load directly, without the need for additional transistors. The drainterminals of MOSFET 108 and MOSFET 110 are connected together to form anoutput node. Feedback resistor 123 and the audio load are connected tothis node. FIG. 22 shows the connection of resistor 400 and resistor 401in that version of the circuit. The circuit of FIG. 22 can also beconnected as FIG. 21 to form an output node, and drive aground-referenced load. The same holds true for the operation of thenegative half of the circuit.

The current in transistor 108 is therefore also fully defined by thevoltages across source resistors 114 and 118 and their counterparts 116and 120. Since the voltage across resistors 114 and 118 and theircounterparts 116 and 120 are temperature stable due to the action of thestable reference, the current in transistor 108 is stabilized. Thecircuit is modulated by driving a voltage from the output of thepreceding stage into the inverting input of the circuit of FIG. 7a atresistor 122. A voltage feedback loop from the output node of the output(current gain) stage is provided through resistor 123. The ratio of thevalues of resistor 123 to resistor 122 is on the order of from about10:1 to about 50:1. In an actual embodiment of the present inventionresistor 122 was selected to be 1.5 KΩ and resistor 123 was selected tobe 39 KΩ.

FIG. 7b is a schematic diagram of an alternate MOSFET voltage gain/phasesplitter stage suitable for use in the present invention. In the circuitof FIG. 7b, the modulating voltage is driven into the non-invertinginputs of the operational amplifiers 92 and 98, while the reference isapplied through the inverting inputs of the operational amplifiers 92and 98. Those of ordinary skill in the art will recognize that thisarrangement causes the overall circuit to become non-inverting.

In the embodiment of FIG. 7b, the operational amplifier 92 drivingMOSFET 94 compares the voltage across resistor 96 with the referencevoltage. The operational amplifier 92 will cause the MOSFET 94 to drivecurrent into the resistor 96 until the voltages at the inverting inputand non-inverting input of operational amplifier 92 are equal, at whichtime the circuit will stabilize. The current in the MOSFET transistor 94is therefore defined and stabilized. The resulting voltage across sourceresistor 118 is also fully defined by the same current and the selectedvalue of the resistor 96. The negative half of the circuit operates inthe same manner as just described for the positive half

The current in MOSFET transistor 108 is therefore also fully defined bythe voltages across resistors 114 and 118. Since the voltage acrossresistor 118 is temperature stable due to the action of the stablereference, the current in transistor 108 is stabilized. The modulatingvoltage is driven into the non-inverting inputs of the operationalamplifiers 92 and 98, while the reference voltage is applied through theinverting inputs of the operational amplifiers 92 and 98. Thisconnection allows the overall circuit to become non-inverting.

Referring now to FIG. 8, a schematic diagram of the positive half of anoutput, or current gain stage 16 of the amplifier 10 of FIG. 2 ispresented. FIG. 8 illustrates the basic circuit and quiescent biasingtechnique without the additional circuitry required to modulate theoutput current.

MOSFET transistor 130 is connected in a feedback loop with operationalamplifier 132. The circuit is arranged so that the operational amplifier132 drives the gate of the transistor 130 with a voltage, which causescurrent to flow in the transistor. Resistor 134 converts the resultingcurrent to a voltage, which is fed to the inverting input of operationalamplifier 132. A stable reference voltage derived from a referencesource 136, such as an LM385 2.5 integrated circuit, is applied to thenon-inverting input of operational amplifier 132 using an appropriateresistor network as in the connection shown. More specifically, theresistor network comprises resistors 138, 140, 142, and 144, which serveto divide the reference voltage and allow for adjustment to a slightlydifferent value for each input of the operational amplifier 132 in orderto establish a nominal quiescent current flow in the output transistors,typically in the range of from about 20 mA to about 30 mA. Resistor 146is used to limit input current to the non-inverting input of operationalamplifier 132, and to provide a voltage compliant node which is isolatedto some degree from the reference divider formed by resistors 138 and140.

Resistor 148 serves to control the effective open loop gain of theoverall feedback loop of operational amplifier 132, which includes theextremely high impedance gate/source path of MOSFET 130. This connectionmay contribute to stability with certain operational amplifiers.Similarly, a small value capacitor of about 1,000 pF may be used inparallel with, or in place of, resistor 148. A diode with its anode atthe inverting input and its cathode at the output of operationalamplifier 132 may also be used to limit the negative-going voltage swingof operational amplifier 132. These techniques are all known to those ofordinary skill in the art and may be applied as needed to control thebehavior of the circuit during turn on and turn off transitions.

Under quiescent conditions, the current flowing in the output MOSFETtransistor 130 is determined by the operational amplifier 132, whichcompares the voltage in the output sense resistor 134 with the voltagefrom the stable references. When the two inputs of operational amplifier132 are at the same voltage, the circuit is in a stable state.

Initially, there is no current flowing in the MOSFET 130. This causesthe inverting input of the operational amplifier 132 to be at a lowervoltage than its non-inverting input because there is no current flowingin the sense resistor 134, and therefore no voltage is being generatedas a result. The values of resistors 138, 140, 142, and 144 are selectedto ensure this initial condition, or some form of adjustment may beprovided to create it, such as a variable resistor string placed betweenthe two nodes in the resistor divider biasing the non-inverting inputsof operational amplifier 132 and its counterpart in the negative half ofthe output stage. When the required condition of voltages exists, theoperational amplifier 132 will cause its output.to slew in the positivedirection, driving a positive voltage to the gate of the power MOSFET130, commanding current to flow.

As current flows, the voltage developed across the sense resistor 134 isapplied to the inverting input of the operational amplifier 132 throughan appropriate resistor network, typically having a voltage division ofabout one half. As this voltage approaches the magnitude of the voltageapplied to the non-inverting input, the output voltage of operationalamplifier 132 stabilizes, thereby causing the current in the MOSFET tostabilize. As this takes place, the voltage across the sense resistor134, which is fed back to the operational amplifier 132, will in turnstabilize and the circuit will remain in a stable, quiescent conditionuntil some modulation is applied to one of the voltages applied toeither of the inputs to operational amplifier 132, with such modulationbeing made possible by the presence of the resistor strings.

Temperature stability of current flow in the circuit is assured becausecurrent in the MOSFET 130 is represented by a voltage across the senseresistor 134, which is in turn compared to a stable reference voltagederived from voltage reference 136. In the case where multiple MOSFEToutput transistors are placed in parallel, they would each be providedwith a sense resistor, and the separate voltages compared in separateoperational amplifiers, but since they are all referenced to the samestable voltage, each MOSFET output transistor is will be commanded todrive currents that will match to within a few percent of each other,and which would be dependent on the tolerances of the components used.Since 1% tolerance resistors are in common use, the circuit can have 1%matching of the currents in paralleled MOSFET output transistors simplyby building it with readily available parts.

Once the connection and quiescent condition described is established,the current in the MOSFET output transistor is modulated by applying anappropriate voltage, typically an audio waveform, to one of the resistorstrings connected to one of the inputs of operational amplifier 132.Therefore, it is possible to drive the current for conditions when largecurrents must be driven into an audio load.

Introducing a modulating signal in order to drive an audio signal poweroutput can be done in several ways. Illustrative ones of those ways aredepicted in FIGS. 9 through 16, to which attention is now drawn.

Referring first to FIG. 9, which comprises the leftmost portion of thecircuit shown in FIG. 8, the resistance of resistor 138 of FIG. 8 may bedivided into two component resistors 150 and 152, thus creating a point154 that can be modulated with a voltage. A small signal common emitterNPN bipolar transistor 158, having its base connected to a source of amodulating signal, its emitter connected to the output node 156, and itscollector connected to point 154 will modulate the feedback voltage fromresistor 144, causing the operational amplifier 132 to command currentflow in the MOSFET output transistor 130 of FIG. 8. This in turn willcause a voltage slewing condition of the output node 156. The bipolartransistors in this connection may be biased to a partially conductingstate using traditional techniques in order to emulate the action of atraditional bi-polar class AB output stage.

The circuit of FIG. 9 has the disadvantage that current gain is inherentin the bipolar transistor 158. This gain can be quite high, and is alsonon-linear, being exponential over several decades of collector current,as is well understood in the art. This will generate a non-linearmodulating signal which can introduce voltage gain into this stage ofthe power amplifier creating a condition which is un-desirable forcircuit stability.

Referring now to FIG. 10 a circuit is shown wherein the bipolartransistor 156 is placed in the feedback loop of an operationalamplifier 160. In this connection, the collector current is forced tobecome a linear analogy of the voltage across resistor 162. Resistor 164provides a compliant connection in the event that bipolar transistor 156becomes saturated.

This circuit avoids the condition of voltage gain in the output stage ofthe amplifier, since the voltage changes in the resistor network biasingthe operational amplifier 160 will also be forced to be linear.

It would be a distinct advantage to have a current limiting functionbuilt in to the output stage of the amplifier. To do this, it isnecessary only to define the limits of voltage modulation applied to theoperational amplifier 160. It is also considered desirable to avoidapplying currents or voltages to the inverting input of the operationalamplifier 160 other than those resulting from the feedback and referenceconnections, or to introduce a connection which would vary the impedanceseen by the inverting input.

These conditions can be met by applying a modulating voltage only to theresistor network connected to the non-inverting input of the operationalamplifier 132 comprising resistors 140 and 142. Further, by defining theminimum and maximum voltage levels applied at the non-inverting input ofoperational amplifier 132, the minimum and maximum current in eachoutput transistor may be also be fully defined.

The circuit of FIG. 11 can accomplish this. In the circuit of FIG. 11,resistor 140 has been replaced by series connected resistors 170 and172. PNP bipolar transistor 174 has its emitter connected to the top ofresistor 170 and its collector connected to the bottom of resistor 170.The base of PNP transistor 174 is driven from NPN transistor 176 throughits collector resistor 178. The modulating signal is applied to the baseof NPN transistor 176.

When both transistors shown are in a non-conducting state, the resistorstring 142, 170 and 172 causes the circuit to establish a quiescentoperating point. When PNP transistor 174 is fully conducting, resistor170 is short circuited out of the circuit, and current flows in theoutput transistor, the magnitude of which is defined by the resultingmodulating voltage, which is in turn fully defined by the resistorvalues in the bias string connected to the non-inverting input throughresistor 146.

To eliminate the non-linear gain of the bipolar transistors in thecircuit of FIG. 11, the circuit of FIG. 12 may be employed. In thisembodiment, resistor 140 is replaced by two series connected resistors180 and 182. An operational amplifier 184 has the audio input signalapplied to its non-inverting input. A diode 186 having its anodeconnected to the output of operational amplifier 184 and its cathodeconnected to the inverting input of operational amplifier 184. A diode188 has its cathode connected to the output of operational amplifier 184and its anode connected to one end of a feedback resistor 190. The otherend of feedback resistor 190 is connected to the inverting input ofoperational amplifier 184. Amplifier 184 is configured as a precisionrectifier. This circuit is known in the art to remove the diode forwardvoltage drop from signal rectifier circuits. In this embodiment,positive going signals are applied to the junction of resistors 180 and182 and affect the voltage applied to the non-inverting input ofoperational amplifier 132 through resistor 138. Complementary actionoccurs during negative voltage swings.

Resistor 190 has a value chosen to be many times larger than theimpedance of the divider string comprised of resistor 180 and 182 tominimize the loading of this point. As the circuit drive swingsnegative, the diodes in the positive half of the circuit become reversebiased, leaving the transistor 130 with only the previously definedquiescent current flowing.

FIG. 13 illustrates a circuit employing a small signal MOSFET 190 as avoltage variable resistor. Resistor 140 is replaced by series connectedresistors 192 and 194. The MOSFET 190 shunts resistor 192 in the biasingstring connected to the non-inverting input of the operational amplifier132. An operational amplifier 196 drives MOSFET 190. The audio inputsignal is applied to the non-inverting input of operational amplifier196 and a feedback resistor 198, its value chosen to provide sufficientfeedback current to operational amplifier 196 to minimize voltage errorsdue to the amplifier input bias current, is connected between the sourceof MOSFET 190 and the inverting input of operational amplifier 196.

The circuit of FIG. 13 allows the string to perform its function in themost pure sense, since the MOSFET gate isolation is high enough so thatno currents or offsets are injected into the resistor string as a resultof leakages or parasitic connections, as is the case with many of theother connections shown. Minimum and maximum currents are defined solelyby the resistor values selected, and interference with the operationalamplifier operating parameters is avoided. The circuit of FIG. 13 mayalso be realized with a junction FET of either N-channel or P-channel,or an FET device designed to be used as a variable resistor.

FIG. 14 is a schematic diagram of the positive half of a circuit forapplying the audio modulation voltage which requires only one additionalreference diode such as the LM385z. The outputs of the circuit of FIG.7b are connected to the reference diode 200, which is in turn connectedto the reference resistive divider string comprising resistors 140 and142, associated with the non-inverting input of operational amplifier132 through the signal diode 202. This diode 202 allows the audiodriving voltage to disconnect from the reference divider string duringexcursions of negative drive voltages. This allows the un-driven half ofthe circuit to be quiescent. The negative half of the circuit is amirror image of the positive half shown in FIG. 14 and contains a signaldiode like 202 having its cathode, rather than its anode, connected tothe reference divider resistor string in the negative portion of thecircuit.

FIG. 15 is a schematic diagram of a simplified circuit that retains thestability of previous circuits yet is economical. The reference diode136, and the corresponding reference diode in the negative half of thecircuit, are both replaced by a single reference diode 136′. Bothresistor dividers, comprised of resistors 140 and 142, and 150, 152, and144 of FIG. 9 are connected to this reference point. This referencevoltage is set to be about 2.5 volts.

The stable reference is then modulated by the outputs of FIG. 7b, or 7a. The diode 136′ is placed so that as the modulating voltage movespositive, the voltage at the junction of resistor 150 and resistor 152is clamped to the output reference point, causing the voltage at thenon-inverting input of 132 to be offset in the positive direction,thereby causing current to flow in transistor 130, as previouslydiscussed.

In the circuit of FIG. 15, resistor 148 and its counterpart in thenegative half of the circuit may both be replaced by a signal diode 204as shown to clamp the output of amplifier 132. In this connection, theMOSFET transistor in the complimentary half of the circuit is allowed toturn completely off during voltage modulation, as opposed to the otherdisclosed embodiments which leave the quiescent current flowing duringvoltage modulation.

FIG. 16 is a schematic diagram illustrating the driving of multipleMOSFET transistors connected in parallel, including the components thatmust be duplicated, and those that are shared by all sections of thecircuit. As may be seen from an examination of FIG. 16, the circuitshown therein is largely a repetition of the circuit of FIG. 8, and isshown driving two MOSFET output devices. A common reference voltagesource 136 is used for both MOSFETS, as are resistors 140 and 142.Components which must be duplicated for each MOSFET output deviceemployed are MOSFET 130, operational amplifier 132, and resistors 134,138, 144, 146, as indicated by MOSFET 130′, operational amplifier 132′,and resistors 134′, 138′, 144′, 146′.

Current feedback is becoming known to reduce audible distortion inindividual loudspeakers. These circuits typically use a small valuesense resistor having a value of about 0.1 ohm referenced to ground.

Amplifier circuits employing current feedback stages have a certaindisadvantage in that a frequency response curve is imposed on theloudspeaker which is an analogy of the loudspeaker impedance. Such aresponse curve is illustrated in FIG. 17, a graph showing speaker voicecoil impedance vs. frequency for typical loudspeakers as measured infree air. This aspect of current feedback tends to limit theapplicability of current feedback to complete systems for which they areprimarily designed.

In some cases when it is desirable to avoid having the speakers act asIf they are altered, such as when driving a complete speaker system witha crossover network connecting multiple speaker drivers, and which wasdesigned to be used without current feedback, the circuit of FIG. 18 maybe usefully employed.

FIG. 18 is a schematic diagram of an audio amplifier 210 according tothe present invention including an improved circuit for developing avoltage which may be applied to a current feedback amplifier.

Those of ordinary skill in the art will recognize that the amplifier 210of FIG. 18 is substantially similar to the circuit earlier describedherein with reference to FIG. 2. Elements of the amplifier 210 of FIG.18 which are the same as corresponding elements of the amplifier 10 ofFIG. 2 will be indicated by the same reference numerals as theircounterparts in FIG. 2.

As in the amplifier of FIG. 2, the first stage of amplifier 210comprises a voltage feedback amplifier 12. While in the amplifier 10 ofFIG. 2, the output of voltage feedback amplifier 12 is presented tovoltage gain/phase splitter stage 14, the output of voltage feedbackamplifier 12 of FIG. 18 is presented to a current feedback stagecomprising an operational amplifier 212 having its inverting inputbiased by resistors 214, 216, and 218. Operational amplifier 212 isconnected as a current feedback amplifier. Resistor 214 comprises asignal input resistor, resistor 216 comprises a feedback resistor, andresistor 218 a gain setting resistor, which acts only to set the gain ofcurrent feedback signals applied to the non inverting input as is knownin the art.

Current feedback is supplied to the non-inverting input of operationalamplifier 212. Transformer 220 is employed to level translate thefeedback signal. The primary of transformer 220 is placed in series withthe output of the amplifier 212 and loudspeaker 222 across senseresistor 224. Rather than being connected to ground, sense resistor 224is located between the amplifier output node and the loudspeaker and isused to develop the voltage for the primary winding of transformer 220.The secondary winding of transformer 222 is connected between thenon-inverting input of operational amplifier 212 and ground. Resistor226 is used to terminate the transformer primary winding, and capacitor228 is employed to control bandwidth.

In a typical arrangement of the current feedback circuit of FIG. 18, theturns ratio of transformer 220 and the gain set by resistors 214, 216,and 218 are selected such that a gain within a range of from about 0.5to about 2 is applied to the input audio signal, and a gain in the rangeof about 10 to about 100 is applied to the current feedback voltage. Inactual operating embodiments of the present invention, resistors 214,216, and 218 were selected to have values of 16 KΩ, 12 KΩ, and between1.2 KΩ-3 KΩ respectively. Those of ordinary skill in the art willreadily be able to configure such a circuit for variations of theoperating conditions disclosed herein.

Transformer 220 is phased such that either degenerative or regenerativefeedback is obtained, with degenerative feedback being considereddesirable for reduction of audible distortion and regenerative feedbackbeing applicable in certain specialty applications such as guitaramplifiers. Those of ordinary skill in the art will appreciate that ifregenerative feedback is employed, care must be taken to prevent thecircuit from oscillating uncontrollably. Resistor 226 and capacitor 228are selected such that the transformer is terminated into a matchedresistive load for the particular transformer used and that thebandwidth of the feedback signal is limited to below about 30 KHz suchthat overall stable operation of the circuit is assured.

The output of the current feedback circuit drives voltage gain/phasesplitter stage 14. As in amplifier 10 of FIG. 2, voltage gain/phasesplitter stage 14 drives current drive stage 16 through bias adjustmentstage 18. Current drive stage 16 drives an output stage 20 comprisingparalleled individual output devices.

The circuit of FIG. 18 has certain advantages when the goal of employingcurrent feedback is to cause the loudspeaker, through action of thepower amplifier, to behave as if certain of its physical parameters havebeen altered.

By locating the sense resistor 224 between the output stage 20 andloudspeaker 222, and applying a feedback connection as shown, flatfrequency response can be achieved. Since the current flowing in theloudspeaker system 222 must also flow In resistor 224 there will be avoltage developed across that resistor that represents the currentflowing in the loudspeaker system, and which is useful in reducingaudible distortion.

Those of ordinary skill in the art will recognize that the transformerconnection disclosed in FIG. 18 may be replaced with an operationalamplifier, configured as an ordinary differential amplifier to achievethe function of level shifting and common mode rejection required toapply the error signal developed across sense resistor 224 to thecurrent feedback operational amplifier 212.

Those of ordinary skill In the art will recognize that operationalamplifier 132 and its counterparts included in the disclosed poweramplifier topology will require a bias voltage supply that is referencedto the output voltage point of the amplifier and that the reference, orcenter point, of this supply is required to float with respect to systemground, and to be driven to the limits of voltage excursion required bythe audio output signal. As is known in the art, such a bias voltagesource may also be configured as a temperature stable voltage source,which can therefore additionally perform the function of the stablereference 136 as referred to in this disclosure.

Such a bias supply may be realized in several ways familiar to those ofordinary skill in the art, two of which are illustrated in FIGS. 19a and19 b.

Referring now to FIGS. 19a and 19 b, two schematic diagrams illustratecircuits for supplying bias voltages to the amplifier circuits disclosedherein. The bias supply circuit 230 of FIG. 19a includes a positive halfcomprising a resistor 232 in series with diode 234 and zener diode 236all connected between the positive voltage rail V++ and the output node238 of the audio amplifier. A filter capacitor 240 is connected inparallel with the zener diode 236. A negative half of the bias supplycircuit 230 comprises a resistor 242 in series with diode 244 and zenerdiode 246 all connected between the negative voltage rail V++ and theoutput node 238 of the audio amplifier. A filter capacitor 248 isconnected in parallel with the zener diode 246.

The bias supply circuit 250 illustrated in FIG. 19b utilizes a centertapped power transformer having the center tap connected to the outputnode 238 of the audio amplifier. A first rectifier diode 254 has itsanode connected to one end of the secondary winding of transformer 252and its cathode connected to positive output node 256. A filtercapacitor 258 is connected between positive output node 256 and outputnode 238 of the audio amplifier. A second rectifier diode 260 has itscathode connected to the other end of the secondary winding oftransformer 252 and its anode connected to negative output node 256. Afilter capacitor 264 is connected between negative output node 262 andoutput node 238 of the audio amplifier.

While embodiments and applications of this invention have been shown anddescribed, it would be apparent to those skilled in the art that manymore modifications than mentioned above are possible without departingfrom the inventive concepts herein. The invention, therefore, is not tobe restricted except in the spirit of the appended claims.

What is claimed is:
 1. A phase-splitter/level shifter comprising: asingle-ended input node; a feedback node coupled to ground through ashunt resistor; first and second complimentary output nodes; first andsecond complimentary power supply rails; a first load resistor coupledbetween said first power supply rail and said first output node; asecond load resistor coupled between said second power supply rail andsaid second output node; a first MOS transistor having a drain coupledto said first output node, a source coupled to said feedback nodethrough a source resistor and a gate; a second MOS transistor having adrain coupled to said second output node, a source coupled to saidfeedback node through a source resistor and a gate; first and secondcomplimentary reference current sources; a first operational amplifierhaving a non-inverting input coupled to said single-ended input node, aninverting input coupled to said second reference current source, and anoutput coupled to said gate of said first MOS transistor; a secondoperational amplifier having a non-inverting input coupled to saidsingle-ended input node, an inverting input coupled to said firstreference current source, and an output coupled to said gate of saidsecond MOS transistor; and a variable resistor coupled between saidsource of said first MOS transistor and said second MOS transistor.